Method and device for compensation of unbalance in a series compensated converter station

ABSTRACT

A series-compensated converter station included in an installation for transmission of high-voltage direct current comprises a converter (SR1, SR2) with at least one 6-pulse bridge (BR). Via series capacitors (SCR, SCS, SCT) the 6-pulse bridge is connected to a three-phase alternating-voltage network (N1, N2) with a fundamental frequency (f 01 , f 02 ). Control equipment (CE1, CE2) generates an ordered value (AOL) of a control angle (α) for valves (V1-V6) included in the 6-pulse bridge in dependence on a limiting signal (AMAXL) capable of being influenced. An amplitude signal (AMPL) is formed which corresponds to the amplitude (C 1 ) for a component (C 1  cos(2πf 0  t+φ 1 )) of the fundamental frequency in the direct voltage (Udb) of the 6-pulse bridge and a compensating signal (ACOMP) is continuously calculated in dependence on a sensed voltage (Un1, Un2) at the converter station and on the amplitude signal. The limiting signal is formed in dependence on the compensating signal for the purpose of maintaining the commutating margin (γ m ) for the valves equal to at least the preselected value (γ p ) in case of unbalance between the voltages (UCR, UCS, UCT) of the series capacitors. (FIG. 6)

TECHNICAL FIELD

The present invention relates to a method for control of aseries-compensated converter station, included in an installation fortransmission of high-voltage direct current, and to a device forcarrying out the method.

By a series-compensated converter station is meant, in this connection,a converter station, the converter bridges of which are connected to analternating-voltage network via series capacitors, possibly with anintermediate transformer.

BACKGROUND OF THE INVENTION

An installation for transmission of high-voltage direct current betweentwo alternating-voltage networks comprises two converter stations, eachconnected on its a.c. side to a separate one of the alternating-voltagenetworks, and a common d.c. connection. The d.c. connection may be inthe form of an overhead line and/or cable and may also in certain partsconsist of ground or water instead of a metallic conductor. Each of theconverter stations comprises a converter, series capacitors, usually atleast one converter transformer, as well as shunt filters for generationof reactive power and filtering of harmonics. The converters arenormally line-commutated, current-source converters, by which it is tobe understood that the current commutation between the valves of theconverters takes place by means of voltages occurring in thealternating-voltage network, and that the d.c. connection, viewed fromthe converters, occurs as a stiff current source.

For the purpose of reducing the harmonics generated by the converters,especially the 5th and 7th harmonics, each of the converters usuallyconsists of two mutually series-connected six-pulse bridges, theconverter transformer being provided with two secondary windings with amutual phase shift of 30°. Each of the converter bridges is connected tothe alternating-voltage network via series capacitors and a separatesecondary winding on the converter transformer.

During normal operation, one of the converters, hereinafter referred toas the rectifier, operates in rectifier operation, and the other,hereinafter referred to as the inverter, operates in inverter operation.Control equipment for the respective converter generates a controlsignal corresponding to a control angle α at which firing pulses areapplied to the valves of the converters.

For the purpose of minimizing the consumption of reactive power by theconverters, and reducing the stresses on components included in theconverter stations, it is advantageous to control the rectifier with thesmallest possible control angle α and to control the inverter with acontrol angle which results in the smallest possible extinction angle γ(margin of commutation) without jeopardizing the controlled operation.The control system of the installation is, therefore, usually designedsuch that the inverter is controlled to a suitable maximum directvoltage for the operating conditions of the installation, taking intoconsideration safety margins with respect to commutating errors, voltagevariations on the a.c. network, and other deviations from nominaloperation which may occur whereas the rectifier is controlled by currentcontrol. The reference value of the current control is formed independence on a current order, which in turn is formed in dependence ona power order and the prevailing direct voltage in such a way that thedirect current and hence the transferred active power remain at adesired value.

Usually, the control equipment for rectifiers and inverters is designedidentically, whereby in the rectifier a current controller is activatedand in the inverter control equipment for a control with the aim ofmaintaining the extinction angle at, but not lower than, a preselectedlowest value is activated. Between the control angle α, the extinctionangle γ, and the overlap angle u, the known relationship α+u+γ=180°prevails. The control equipment of the inverter is usually designed suchthat its control angle is formed in dependence on a limiting signal.Such a limiting signal may, in turn, be formed in dependence on sensedvalues of direct current and available commutating voltage, independence on predicted values of the extinction angle, or by means of afeedback control of the extinction angle.

The control system of the converters is usually designed to generatefiring pulses to the respective valves with mutually identicalintervals, so-called equidistant control.

With series compensation, several advantages are obtained. The seriescapacitors are charged periodically by the current which flows throughthem and the voltage across the capacitors thus generated provide anaddition to the commutating voltage across the valves of the converter.The commutating voltage becomes phase-shifted relative to the voltagesof the alternating-voltage network in such a way that, with control andextinction angles still related to the phase position for the voltagesof the alternating-voltage network, the valves in rectifier operationmay be controlled with control angles smaller than zero and in inverteroperation with extinction angles smaller than zero (although thecommutating margin, related to the commutating voltage of the valve, is,of course, greater than zero). This makes possible a reduction of thereactive power consumption for the converters. In this way, the need ofgeneration of reactive power in the shunt filters is reduced, and theshunt filters may thus be dimensioned substantially on the basis of theneed of harmonic filtering.

The charging current of the capacitors and hence the voltage thereof areproportional to the direct current in the d.c. connection, and bysuitable dimensioning of the capacitors, the dependence of the overlapangle on the magnitude of the direct current may be compensated for.This means that the series compensation contributes to maintaining thecommutating margin of the valves also in case of fast currenttransients. The control of the inverter means that the inverter, atleast without series compensation, exhibits a negative current/voltagecharacteristic, which, especially in those cases where the d.c.connection comprises a long cable, in case of voltage reductions in thealternating-voltage network may lead to an avalanche-like growth of thecurrent. The series compensation influences the current/voltagecharacteristic of the inverter in a stabilizing direction, and by asuitable choice of series capacitors it may also be brought to bepositive.

For a general description of the technique for transmission ofhigh-voltage direct current, reference is made to Erich Uhlmann: PowerTransmission by Direct Current, Springer Verlag, Berlin Heidelberg NewYork 1975.

A general description of the mode of operation of the converter stationwith series capacitors introduced into the ac connections between theconverter transformer and a converter in a six-pulse bridge connectionis given in John Reeve, John A. Baron, and G. A. Hanley: A TechnicalAssessment of Artificial Commutation of HVDC Converters with SeriesCompensation (IEEE Trans. on Power Apparatus and Systems, Vol. PAS-87,October 1968, pages 1830-1840).

However, series compensation of the converter station means that thecommutating voltage of the valves is dependent on both amplitude andphase for the current-dependent voltage across the respective seriescapacitor. During stationary undisturbed operation with symmetricalphase currents, the mean value of the voltage across the respectivecapacitor remains equal to zero and the capacitor voltages are identicalbetween themselves. Their contribution to the commutating voltage thusremains equal for all the valves included in the converter bridge. Incase of fast changes in direct current and/or control angle for theconverter bridges, however, capacitors in different phases will becharged with a different current-time area and hence assume differentvoltages. An unbalance in capacitor voltage and hence in commutatingvoltage thus arises between the different phases.

When the operating state returns to stationary undisturbed operation,the unbalance is reduced by itself in that it entails different overlapangles and conduction intervals for different valves. This process,however, occurs relatively slowly, especially during operation with lowcurrent and may typically take several seconds.

The unbalance also means that the voltage across the direct-voltageterminals of the converter bridges will contain an alternating-voltagecomponent of the same frequency as the fundamental-tone frequency of theconnected alternating-voltage network, usually 50 or 60 Hz. A method anda device for damping oscillations in a power transmission system at ornear fundamental frequency are described in the patent specification WO94/07291.

However, an unbalance in the capacitor voltages also means that thecommutating margin is reduced for certain valves in the converterbridge, which in turn entails an increased risk of commutating errors.This increased risk remains, although to a decreasing extent, alsoduring the time the unbalance is reduced.

SUMMARY OF THE INVENTION

The object of the invention is to provide a method of the kind describedin the introduction, which, in case of unbalances occurring in thecapacitor voltages, influences the ordered control angle such that acommutating margin is obtained which is not lower than that which isaimed at, and a device for carrying out the method.

What characterizes a method and a device according to the invention willbecome clear from the appended claims.

Advantageous improvements of the invention will become clear from thefollowing description and claims.

By means of the invention, the advantages of the series compensation areutilized without the risk of commutating errors increasing in connectionwith transients in direct current and/or control angle. For the inverterthis is of importance since the aim of its control equipment duringnormal operation is that the commutating margin should remain at but notbe below a preselected lowest value. For the rectifier, this isimportant above all in case of rapid control angle changes, for exampleduring such fault conditions in the dc connection where the rectifier isto be rapidly controlled towards the greatest possible negative voltage,which is achieved by ordering the control angle to a value near 180°.Such control angle changes means, on the one hand, that the risk ofunbalance of the kind mentioned increases and, on the other hand, thatthe rectifier will operate near its commutating margin.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be explained in greater detail by description ofembodiments with reference to the accompanying drawings, wherein

FIG. 1 schematically shows an installation for transmission ofhigh-voltage direct current with series-compensated converter stations,

FIG. 2 shows a converter bridge connected via series capacitors to athree-phase alternating-voltage network,

FIG. 3 shows two series-connected converter bridges according to FIG. 2,

FIG. 4 shows in the form of a block diagram parts of control equipmentfor the converter stations according to FIG. 1, in one embodiment of theinvention,

FIG. 5 shows in the form of a block diagram an embodiment of a currentcontroller for control equipment according to FIG. 4,

FIG. 6 shows in the form of a block diagram a limitation of the maximumcontrol angle in inverter operation for control equipment according toFIG. 4 in one embodiment of the invention,

FIG. 7 shows in the form of a block diagram the formation of anamplitude signal and a damping signal in one embodiment of theinvention,

FIG. 8 shows in the form of a detailed block diagram one way of formingan amplitude signal in an embodiment of the invention according to FIG.7,

FIG. 9 shows in the form of a block diagram the formation of anamplitude signal and a damping signal in another embodiment of theinvention,

FIG. 10 shows in the form of a detailed block diagram one way of formingan amplitude signal in an embodiment of the invention according to FIG.9,

FIG. 11 shows in the form of a detailed block diagram one way of forminga damping signal in an embodiment of the invention according to FIG. 9,

FIG. 12 shows in the form of a block diagram one way of forming adetection signal for detection of a fundamental frequency oscillation inan embodiment of the invention according to FIG. 7, and

FIG. 13 shows in the form of a block diagram one way of forming adetection signal for detection of a fundamental frequency oscillation inan embodiment of the invention according to FIG. 9.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The following description relates both to the method and to the device,and the figures can thus be regarded both as signal flow diagrams andblock diagrams of devices. The expressions "calculating value","(calculated) value" and "signal" are therefore used synonymously.

It is assumed in the following that the control angle α, the angle whencommutation is started, and the extinction angle γ, the angle whencommutation is terminated, are related to the voltages of the respectivealternating-voltage network in a conventional manner. By commutatingmargin γm is meant the extinction angle related to the commutatingvoltage across a valve in a converter bridge. For anon-series-compensated converter station, the extinction angle γ is thusequal to the commutating margin γm, whereas for a series-compensatedconverter station the extinction angle γ generally deviates from thecommutating margin γm and may also assume values less than zero.

FIG. 1 shows an installation for transmission of high-voltage directcurrent between two three-phase alternating-voltage network N1 and N2,only roughly indicated. Each one of the alternating-voltage networks hasa fundamental frequency f₀₁ and f₀₂, respectively, usually 50 or 60 Hz.

A converter SR1 is connected with its alternating-voltage terminals tothe network N1 via series capacitors SC1 and a transformer T1 and aconverter SR2 is connected with its alternating-voltage terminals to thenetwork N2 via series capacitors SC2 and a transformer T2. Each one ofthe transformers is equipped with a tap-changer TC1, TC2, respectively,marked with an arrow in the figure. A d.c. connection L1, L2 connectsthe direct-voltage terminals of the converter SR1 to the correspondingdirect-voltage terminals on the converter SR2. The impedances of thed.c. connection are designated Z1, Z2, respectively. Further, shuntfilters (not shown in the figure) for generation of reactive power andfiltering of harmonics are connected to the respectivealternating-voltage network.

For the description of the embodiment it is assumed that the converterSR1 operates as a rectifier and the converter SR2 operates as aninverter, but both converters are adapted to be able to operate in aknown manner both as rectifiers and inverters.

The converters may be designed in a known manner as two series-connected6-pulse bridges to form a 12-pulse connection, whereby each of thetransformers comprises two secondary windings with a mutual phase shiftof 30°, for example one secondary winding in Y connection and onesecondary winding in Δ connection. FIG. 2 shows a six-pulse bridge BR,comprising six mutually identical valves V1-V6, shown as thyristors inthe figure, connected on the alternating-voltage side via seriescapacitors SCR, SCS, SCT to a three-phase network comprising threevoltage generators GR, GS, GT in series connection with three inductorsLR, LS, LT, respectively, which network constitutes an equivalentcircuit for the above-mentioned transformer, shunt filter andalternating-voltage network. The direct voltage across the 6-pulsebridge is designated Udb.

The commutating voltage for a valve is designated UKVm,n, where index mindicates a decommutating and index n a commutating valve. Duringcommutation from the valve V1 to the valve V3, during which process alsothe valve V2 carries the current, the commutating voltageUKV1,3=US-UCS-UR+UCRT. At a transient delay of the time of firing of thevalve V3, the current IR will flow through the capacitor SCR acorrespondingly longer time and the capacitor voltage UCR will have anaddition dUCR corresponding to the added current-time area. If thecommutating voltages during a stationary condition are designatedUKVSm,n, the commutating voltage voltage will thus in this case beUKV1,3=UKVS1,3+dUCR. Assuming a constantly ordered control angle, thecontrol system for the equidistant control, compared with stationaryconditions, will delay the time of firing of the valve V4 by an amountequal to the transient delay of the time of firing of the valve V3. Thismeans that the valve V2 will have its conduction interval extended justas much as the valve V1. With the polarities described in FIG. 2, thecapacitor voltage UCT will have an addition dUCT=-dUCR. An analysis ofthe commutating voltages over a whole fundamental-tone period shows thatthese voltages will describe a fundamental-tone oscillation with thepeak value 2dUCR, which results in a corresponding fundamental toneoscillation in the direct voltage Udb of the bridge. Also thecommutating margins will vary according to the same pattern, where inthis example the valve V2 will have the greatest and the valve V5 thesmallest commutating margin.

FIG. 3 shows two 6-pulse bridges BR and BR', series-connected on theirdirect-voltage sides, of the kind described with reference to FIG. 2.The voltages UR0, US0 and UT0 are each phase-shifted 30° relative to thecorresponding voltages UR0', US0' and UT0', respectively. For therectifier, the voltage across the direct-voltage terminal of therespective bridge is designated Udb11 and Udb12 and for the inverterUdb21 and Udb22. For the direct voltage Ud1 of the rectifier, thus,Ud1=Udb11+Udb12, and for the direct voltage Ud2 of the inverter,Ud2=Udb21+Udb22.

Each converter is equipped with a piece of control equipment CE1, CE2,respectively (FIG. 1). Each of the pieces of control equipment comprisesa control angle unit CAC for forming an ordered value of the controlangle α, which control angle unit will be described in greater detailbelow, units CFC designed known manner for determining the firing momentof the respective valve in dependence on the ordered value of thecontrol angle α, and CPG for generating control pulses CP1 and CP2,respectively, to the valves included in the converters at the firingmoment. From a power control unit POC, the control angle unit CAC issupplied with a reference value for active power, which reference valueis formed in a known manner. The control angle unit may also be suppliedwith other reference values from superordinate control systems, notshown in the figure, for example for control of reactive power exchangewith the alternating-voltage networks.

The direct voltage Udl of the rectifier and the direct voltage Ud2 ofthe inverter are measured by means of voltage-measuring devices UM1,UM2, respectively, which deliver the measured values UD1 and UD2,respectively. In addition, the voltage measuring devices are adapted todeliver measured values UDB11, UDB12 of the bridge voltages Udb11 andUdb12 and measured values UDB21, UDB22, respectively, of the bridgevoltages Udb21 and Udb22 of the inverter. The current Id through thed.c. connection is measured by means of current measuring devices IM1,IM2, respectively, which deliver the measured values ID1 and ID2,respectively. The voltages Unl and Un2, respectively, of thealternating-voltage networks are measured by means of voltage-measuringdevices UMN1 and UMN2, respectively. which deliver the measured valuesUN1 and UN2, respectively. The measured values UN1 and UN2 also containinformation about the fundamental frequency f₀₁ and f₀₂ of therespective alternating-voltage network.

The pieces of control equipment of the converters are supplied with theabove-mentioned measured values of the operating parameters of theinstallation, that is, the control equipment of the rectifier issupplied with measured values for the voltage of the alternating-voltagenetwork, for direct and bridge voltages of the rectifier, and the directcurrent in the dc connection, and the control equipment of the inverteris supplied with corresponding measured values relating to the inverter.In addition, the pieces of control equipment are supplied (in a mannernot shown in the drawings but known) with input signals representing thefundamental frequency of the respective alternating-voltage network andinformation about the position of the tap-changers and a power-directionsignal RECT/INV, the latter signal indicating rectifier operation andinverter operation, respectively, and being determined in dependence onthe power direction requested by the operator of the installation.

In dependence on measured values and input signals supplied to thepieces of control equipment, the pieces of control equipment of therectifier and the inverter generate control pulses CP1 and CP2,respectively, for the valves of the converters and supply these to therespective valve.

The two pieces of control equipment communicate with each other, knownmanner, via a telecommunication link TL for two-way transmission ofinformation about the operating parameters of the converters.

The respective control equipment may also comprise a tap-changer controlunit, not shown in the figure but designed in a known manner, forgeneration of INCREASE/DECREASE impulses for the tap-changers, whichimpulses are supplied to the operating equipment of the tap-changers.

FIG. 4 shows parts of a piece of control equipment for the converterstations according to FIG. 1, in one embodiment of the invention. Thepieces of control equipment are usually designed identically for bothrectifiers and inverters, and therefore in FIG. 4 and the subsequentFIGS. 5-7 and 9-11, indices 1 and 2, respectively, for indicatingquantities relating to a rectifier and an inverter are not indicated.

The power control unit POC comprises a calculating member IOCAL forcalculating a current order IO as the quotient between a power order POfor transferred active power in the d.c. connection and a measured valueUD of the direct voltage Ud in the rectifier. The current order issupplied to a limiting member 1 for limiting the current order independence on the measured value UD of the direct voltage Ud, suppliedto the above-mentioned limiting member. The output signal IOL from thelimiting member 1 is thereafter supplied to a current controller CCcomprised in the control angle unit CAC as reference value for thiscontroller. The current controller is also supplied with a signal SF,which is formed according to a further development of the invention andwhich will be explained in greater detail in the following.

The output signal AO of the current controller is limited to its minimumand its maximum value in a limiting member 2 by means of limitingsignals AMAXL and AMINL, respectively, capable of being influenced. Theoutput signal AOL from the limiting member 2, which thus is an orderedvalue of the control angle α, is supplied to the unit CFC fordetermining the firing moment of the respective valve.

At each of the firing moments for a valve, the unit CFC also generates asynchronization signal SYNC, the function of which will be described inmore detail below.

FIG. 5 shows an embodiment of the current controller CC. A summator 3forms as output signal the difference of the reference value IOL for thedirect current Id and the measured value ID of this current. Thedifference is supplied to a proportional-amplifying member 4 with a gainGP and to a summator 5. The summator 5 is also supplied with apreselected current margin IOM between the rectifier and the inverter,and forms as output signal the difference of the current margin and theoutput signal from the first summator 3. The output signal from thesummator 5 is supplied to an integrating member 6 with the integrationtime constant 1/GI. The integrating member comprises a limiting member 7which limits the output signal from the integrating member to itsmaximum and to its minimum value in dependence on a limiting signalAMAXLI, capable of being influenced, and the limiting signal AMINL,respectively. A summator 8 is supplied with the signal SF and with theoutput signals from the proportional-amplifying member 4 and from theintegrating member, limited by the limiting member 7. The output signalfrom the summator 8 constitutes the output signal AO of the currentcontroller and forms the sum of the signal SF and the output signal fromthe integrating member, reduced by the output signal from theproportional-amplifying member.

The current orders and the current margins for the rectifier and theinverter are synchronized via the telecommunication link TL, but thecoordination may also be performed in other conventional ways.

The current margin IOM is usually equal to zero for the rectifier andfor the inverter it is set at a value different from zero and with sucha sign that the control equipment of the inverter strives to reduce thedirect current controlled by the rectifier. During stationary inverteroperation, the input signal to the integrating member 6 thus consists ofthe current margin, which means that its output signal will assume itsmaximum value limited by the limiting signal AMAXLI. The output signalfrom the proportional-amplifying member 4 is, in equilibrium state,equal to zero or near zero and therefore, if the signal SF istemporarily disregarded, the value of the control angle α ordered by theinverter is determined by the above limiting signal.

FIG. 6 shows how the limiting signals AMAXL and AMAXLI are formed in anadvantageous embodiment of the invention. A control unit ALCAL forms ina known manner, for example in dependence on parameters such as thedirect current Id and/or the ideal no-load direct voltage Udi0, anoutput signals AMARG as a value of the control angle which, at balancedcapacitor voltages, is assumed to give a commutating margin γm with asatisfactory safety with respect to commutating errors. The outputsignal AMARG is supplied to and limited in a limiting member 9 to itsmaximum and its minimum value by means of limiting signals AMAX andAMIN, respectively. The output signal from the limiting member 9 issupplied to a summator 101, which is also supplied with a compensatingsignal ACOMP which is formed according to the invention and which willbe described in more detail below. The output signal from the summator10, which is the difference of the output signal from the limitingmember 9 and the compensating signal ACOMP, constitutes the limitingsignal AMAXL, which is supplied to the limiting member 2 of thecontrol-angle unit CAC. The signal AMAXL is supplied to a summator 102,which is also supplied with a signal SI which is formed according to afurther development of the invention and which will be described in moredetail below. The output signal from the summator 102, which is thedifference of the limiting signal AMAXL and the signal SI, constitutesthe limiting signal AMAXLI, which is supplied to the limiting member 7of the current controller.

The voltage value UDI0 is formed in a known manner as the output signalfrom a rectifier 11 (FIG. 6), which rectifies the measured value UN ofthe voltage Un of the alternating-voltage network, taking intoconsideration the current transformer ratio.

The limiting signal AMINL is formed in a known manner for the respectiverectifier and inverter.

FIG. 7 shows one embodiment of the invention. An amplitude-value formingunit AVU, comprising a frequency-separating member FSD and anabsolute-value forming member ABS, is supplied with a measured value UDBof the direct voltage Udb of a 6-pulse pulse bridge and forms, for asensed component of the fundamental frequency in the direct voltage, forexample caused by the above-described unbalance in capacitor voltages,as output signal an amplitude signal AMPL.

The measured value UDB here designates one of the above-mentionedmeasured values UDB11, UDB12 of the bridge voltages Udb11 and Udb12 ofthe rectifier and the measured values UDB21, UDB22 of the bridgevoltages Udb21 and Udb22 of the inverter.

If the bridge voltage is assumed to be ##EQU1## where k is a naturalnumber 1, 2, 3, . . . ,

A₀ designates the direct-voltage component of the bridge voltage andC_(k) the amplitude of the k'th harmonic tone thereof. C₁ thusdesignates the amplitude of the component of fundamental frequency inthe bridge voltage and φ₁ its phase angle relative to the firing momentof the valves.

The frequency-separating member FSD is adapted to separate, from thesupplied measured value UDB, the component of the fundamental frequencyand be shown as a bandpass filter. It may consist of a bandpass filterdesigned in some known manner, tuned to the fundamental frequency f₀ ofthe alternating-voltage network. Its output signal, designated C_(1m)*cos(2πf₀ t+φ1), constitutes a measured value of the above fundamentalcomponent and is supplied to the absolute-value forming member ABS. Theabsolute-value forming member is adapted to form the amplitude signalAMPL in dependence on the amplitude C_(1m) such that the amplitudesignal AMPL corresponds to the amplitude C₁ of the component of thefundamental frequency in the bridge voltage. FIG. 8 shows an embodimentof the absolute-value forming member, comprising a rectifier 12, whichis supplied with the signal C_(1m) *cos(2πf₀ t+1), and a low-pass filter13 with a time constant T capable of being influenced, which is suppliedwith the output signal from the rectifier and forms as output signal theamplitude signal AMPL. A comparing member 14 is supplied with the outputsignal from the rectifier 12 and the amplitude signal AMPL and controls,in dependence on the comparison, a selector SEL4. Via the selector thetime constant of the low-pass filter is set a zero when the outputsignal from the rectifier is greater than the amplitude signal andotherwise at a value T1. The value T1 may advantageously be selected tocorrespond to a period of the fundamental frequency.

The amplitude signal AMPL is supplied to a selector SEL1 (FIG. 7),controlled by a detector signal FFD. According to a preselectedcriterion, the detector signal indicates the occurrence of a componentof the fundamental frequency in the bridge voltage and embodiments theformation thereof will be described in more detail below. The occurrenceof such a component is, as described above, connected with variations incommutating margin and the object of the invention is now to achieve, independence on the amplitude signal, a compensation of these variationsso as to obtain a commutating margin which is not lower than the valueaimed at.

For the main circuits of the series-compensated converter station ininverter operation, current/voltage equations may be set up in a knownmanner with the control angle α (related to the voltage of thealternating-voltage network), the direct current Id1, the ideal no-loaddirect voltage Udi02 and the commutating margin γ_(m) of the valve asvariables. If in these equations a constant preselected value γ_(p) ofthe commutating margin of the valve is assumed, the control angle α canbe calculated, suitably iteratively, with the direct current and theideal no-load direct voltage as variables. In formal terms, this can beexpressed such that the control angle is a function G of the directcurrent, the ideal no-load direct voltage and the commutating margin,α=G(Id2, Udi02, γ_(p)). In this connection, the voltages UCR, UCS, UCTare implicitly included, which voltages, as mentioned above, arefunctions of the direct current.

However, in case of an unbalance in the capacitor voltages, as mentionedabove, variations in the commutating margin of the valves will arise,and these variations are dependent on the additions dUCR, dUCS, dUCT tothe balanced capacitor voltages. If these additions are generallydesignated ΔUc, for that valve whose commutating voltage is subjected tothe greatest reduction (in the above example the valve V5), a maincircuit relationship may be set up with a control-angle addition Δα, thedirect current Id2, the ideal no-load direct voltage Udi02, thecommutating margin γ_(m) of the valve, and the unbalance voltage ΔUc asvariables. If in these equations a constant preselected value γ_(p) ofthe commutating margin is assumed, the control-angle addition Δα may becalculated, suitably iteratively, with the direct current, the idealno-load direct voltage, and the unbalance voltage of the capacitors asvariables. In formal terms, this may be expressed such that thecontrol-angle addition is a function F0 of the direct current, the idealno-load direct voltage, the commutating margin, and the unbalancevoltage, Δα=F0(Id2, Udi02, γ_(p), ΔUc). The control-angle addition Δa isthus the addition which, at the unbalance voltage ΔUc, must be added tothe control angle which is formed in dependence on the signal AMARGduring operation with balanced capacitor voltages in order for acommutating margin equal to the preselected value γ_(p) to be maintainedon the valve which, because of the unbalance volrage, is given thesmallest commutating margin. As will be clear from the above, theamplitude signal AMPL corresponds to the amplitude C₁ of the componentof fundamental frequency in the bridge voltage and this amplitude, inturn, corresponds to the unbalance voltage ΔUc.

When the detector signal FFD occurs, the output signal SA thereof is setvia the selector at a value equal to the amplitude signal AMPL,otherwise it assumes the value zero. The output signal SA from theselector is supplied to a calculating unit CALC2 (FIG. 6). Thecalculating unit comprises calculating members, adapted to continuouslycalculate a compensating signal ACOMP corresponding to a control-angleaddition according to the relationship H0, according to which thecontrol-angle addition is the function F0 mentioned above. Thecompensating signal calculated and formed by the calculating unit issupplied to the summator 101 and is subtracted therein from that value,formed by the control unit ALCAL and limited by the limiting member 9,of the control angle which under otherwise similar conditions is validfor balanced capacitor voltages. In this way, thus, the control angle,which is formed in dependence on the limiting signal AMAXL (FIG. 4), isreduced by the value Δα, which in turn means that also the valve which,because of the unbalance voltage is given the smallest commutatingmargin, is given a commutating margin which is not lower than thepreselected value γ_(p).

The function F0 is relatively complicated and a study of representativeplants has shown that a satisfactory compensation for the variations inthe commutating margin may be achieved even when the compensating signalACOMP is formed as a calculated expression according to a relationshipwhich approximately imitates the relationship F0 described above. Inthis way, simplified functions for the dependence of the control-angleaddition on direct current, ideal no-load direct voltage, and unbalancevoltage may be assumed, which means that the calculating memberscomprised in the calculating unit CALC2 may be designed simpler andperform the calculation more rapidly. It has proved to be advantageousto continuously supply the calculating unit with a voltage value UDI0 ofthe ideal no-load direct voltage of the converter bridge and to adaptthe calculating unit to calculate, in dependence on supplied values, acompensating signal ACOMP corresponding to a control-angle additionaccording to a relationship H1 of the form ##EQU2## where K1 is anamplification factor.

FIG. 7 further shows an advantageous improvement of the invention,according to which a damping signal UOD is formed, which is supplied tothe current controller CC for faster damping of the fundamentalcomponent in the voltage of the converter bridge and the unbalancevoltages in the series capacitors. A damping signal-forming unit DSU issupplied with the measured value C_(1m) *cos(2πf₀ t+φ₁) of the componentof the fundamental frequency in the voltage of the bridge, theabove-mentioned signal the synchronization signal SYNC, and the value IOof the current order of the d.c. transmission. The unit DSU comprises aphase-shifting member PHS, an amplitude-adapting member AAD, and anabsolute-value forming member ABS of the same kind as that which isincluded in the amplitude-value forming unit. The phase-shifting membercomprises a sampling circuit SAMP3, a shift register SH3 and a holdingcircuit SAH, controlled by the synchronization signal. The shiftregister has a series input and a parallel output, which output may beconnected to optional positions in the shift register by means of anadjustable selector SEL5, symbolized in the figure by a two-way arrow.The sampling circuit forms a sampled value of the measured value C_(1m)*cos(2πf₀ t+φ₁) upon each synchronization signal, and these sampledvalues are supplied to the shift register. At that position in the shiftregister where the signal is taken out, its input signal has becomedelayed corresponding to a phase addition Δφ. The output signal from theshift register is supplied to a holding circuit, the output signal ofwhich consists of the measured value C_(1m) *cos(2πf₀ t+φ₁) delayed bythe phase addition Δφ, that is, C_(1m) *cos(2πf₀ t+φ₁ -Δφ).

A detailed analysis of the voltage conditions in the converter bridge incase of unbalance in the voltages of the capacitors shows that optimumdamping is achieved when the resultant control-angle change which isachieved in dependence on the damping signal is phase-shifted 120°relative to the fundamental component in the bridge voltage. The phaseaddition Δφ in the shift register is therefore ideally to correspond to120°, but delays in the signal processing and the influence on thefiring moment of the valves mean that the phase addition should beselected somewhat lower than 120°. For example, therefore, the shiftregister may be designed so as to permit a setting of the phase delayequal to 120°±n*12°, where n is a number 0, 1, 2, 3, . . . .

The amplitude-adapted member AAD comprises a multiplier 15, afunction-forming member 16, and a limiting member 17. The output signalC_(1m) *cos(2πf₀ t+φ₁ -Δφ) from the phase-shifting member PHS issupplied to the multiplier for multiplication by the output signal IOCfrom the function-forming member 16. The output signal IOC is formed independence on a value of the current order IO, supplied to thefunction-forming member, in such a way that the output signal IOCdecreases with increasing current order. The output signal from themultiplier is supplied to the limiting member 17 which limits theproduct IOC*C_(1m) *cos(2πf₀ t+φ₁ -Δφ) so as not to exceed a value DMAX.The output signal from the limiting member constitutes the dampingsignal UOD. The multiplication of the signal C_(1m) *cos(2πf₀ t+φ₁ -Δφ)by IOC brings about an amplification adaptation of the intervention fromthe damping signal, taking into consideration that, at large currents, asmaller control-angle change is required to achieve a certain change ofthe capacitor voltage.

The damping signal UOD thus consists of a signal of the formUOD=D*cos(2πf₀ t+φ_(a)), where its amplitude D is formed in dependenceon the product of the output signal IOC from the function-forming member16 and the amplitude C₁ of the component of the fundamental frequency inthe bridge voltage and its phase angle φ_(a) consists of the difference(φ₁ -Δφ) of the phase angle φ₁ for the component of fundamentalfrequency in the bridge voltage and the phase addition Δφ.

The value of DMAX may advantageously be chosen to correspond to an angleof about 5°.

For the damping signal to be able to provide both positive and negativeadditions to the ordered value AOL of the control angle α, the outputsignal from the integrating member 6 is limited to a value AMAXLI whichin magnitude is smaller than the limiting value AMAXL which is suppliedto the limiting member 2 (FIG. 5) of the control-angle unit CAC. This isachieved by supplying the damping signal to the absolute-value formingmember comprised in the damping-signal forming unit, the output signalUODA of the absolute-value forming member being formed in dependence onthe amplitude D for the damping signal.

The output signal UODA is supplied to a selector SEL3, which iscontrolled by the detector signal FFD. When the detector signal ispresent, the output signal SI of the selector assumes a value equal tothe output signal UODA; otherwise, it assumes the value zero. The outputsignal SI from the selector is supplied to the summator 102 (FIG. 6),the output signal of which constitutes the limiting signal AMAXLI formedas the difference of the limiting signal AMAXL and the output signal SI.

FIG. 9 shows a general block diagram for another embodiment of theinvention.

The amplitude-value forming unit, which is supplied with the measuredvalue UDB of the component of the fundamental frequency in the bridgevoltage, the synchronization signal SYNC and a measured value of thefundamental frequency f₀ of the respective alternating-voltage network,is shown in a more detailed block diagram in FIG. 10. For a sensedcomponent of fundamental frequency in the direct voltage, for examplecaused by the above-described unbalance in capacitor voltages, theamplitude-value forming unit forms the amplitude signal AMPL and acosine amplitude signal A and a sine amplitude signal B.

The amplitude-value forming unit comprises an oscillator 18 which, independence on the supplied measured value of the fundamental frequencyf₀ of the respective alternating-voltage network and on thesynchronization signal SYNC for the converter which is connected to thisalternating-voltage network, generates a cosine signal SCOS=cos(2πf₀ t)and a sine signal SSIN=sin(2πf₀ t), where `t` stands for the time, suchthat the phase positions for sine and cosine are mutually identical andtheir phase shifts are related to the synchronization signal SYNC. Inthis embodiment, it is assumed for the sake of simplicity that the phaseshift related to the synchronization signal SYNC is zero.

Further, the amplitude-value forming unit comprises a multiplier 19, asampling circuit SAMP1, controlled by the synchronization signal SYNC, ashift register SH1 with a series input and a parallel output, a summator20 and a multiplier 21.

The measured value UDB is supplied to the multiplier 19, in which it ismultipliedby the cosine signal SCOS. The output signal from themultiplier is supplied to the sampling circuit, which forms a samplingvalue of the product UDB*SCOS at each synchronization signal. Thesampling value is supplied to the shift register in which are stored thelatest sampling values to a number corresponding to at least one periodof the signal SCOS. If the number of sampling values which may be storedin the shift register are designated N, it is advantageous in a 6-pulsebridge to choose N to be 6 or to be an integer multiple of 6. The Nlatest sampling values, corresponding to one or several periods of thesignal SCOS, are summed upon each synchronization signal in the summator20, the output signal of which in the multiplier 21 is normalized bymultiplication by a number 2/N.

If the bridge voltage is set at ##EQU3## where i is a natural number 1,2, 3 . . . , the output signal from the multiplier 21 is a cosineamplitude signal A which is a measure of the amplitude A₁ of the cosinecomponent of the fundamental frequency in the bridge voltage.

The amplitude-value forming unit further comprises a multiplier 22, asampling circuit SAMP2, a shift register SH2, a summator 23 and amultiplier 24.

The measured value UDB of the bridge voltage is also supplied to themultiplier 22, in which it is multiplied by the sine signal SSIN. Theoutput signal from the multiplier 22 is then processed in a manneranalogous to that described above in the sampling circuit SAMP2, theshift register SH2, the summator 23 and the multiplier 24. The outputsignal from the multiplier 24 is a sine amplitude signal B which is ameasure of the amplitude B₁ of the sine component of the fundamentalfrequency in the bridge voltage.

The cosine amplitude signal A and the sine amplitude signal B aresupplied to a calculating member CALC1, which in a known mannercalculates the square root from the sum of the squares of A and B.

It is realized that if the bridge voltage is assumed to be ##EQU4## thesignal calculated by the calculating member CALC1 is a measure of theamplitude C₁ of the fundamental component in the bridge voltage andforms the amplitude signal AMPL. Also in this embodiment of theinvention, the amplitude signal is supplied to the selector SEL1, whichfor the sake of clarity is also shown in FIG. 9. The compensating signalACOMP is then formed in the manner described above.

FIG. 9 further shows another embodiment of the further development ofthe invention according to which a damping signal UOD is formed. Thedamping-signal forming unit DSU comprises a phase-shifting member PHS,an amplitude-adapting member AAD, and a signal-synthesizing member SSD,which members are shown in more detail in FIG. 11. Theamplitude-adapting member is of the same kind as that described abovewith reference to FIG. 7, but in this embodiment the multiplier 15 issupplied with the amplitude signal AMPL and the unit delivers as outputsignal a compensated amplitude signal designated C_(a).

The phase-shifting member PHS comprises a phase-angle forming memberCALC3 and a summator 25. The cosine amplitude signal A and the sineamplitude signal B may assume both positive and negative values and areto be regarded as two components, perpendicular to each other, of avector corresponding to the amplitude signal AMPL, which in turn is ameasure of the amplitude C₁ of the fundamental component in the bridgevoltage. The signals A and B are supplied to the phase-angle formingmember, which in a known manner is adapted to form a value which is ameasure of the phase angle φ₁ for this fundamental component. This ismarked in the figure by the phase-angle forming member comprisingcalculating members adapted to calculate a phase angle φ' according tothe expression φ'=arc tan (-B/A) and by sensing the sign of the cosineamplitude signal A and the sine amplitude signal B, respectively, in thefigure marked by sign(A) and sign(B), respectively, to the value of theprincipal value of the arctan function, which value by definition liesbetween -90° and +90°, correctly taking into consideration the signs ofthe signals A and B and, where applicable, adding an angle 180°.

The value of the phase angle φ₁, formed by the phase-angle formingmember, and a phase addition Δφ are supplied to the summator 25 which asoutput signal forms a compensated phase angle φ_(a) as the difference(φ₁ -Δφ) of the phase angle φ₁ for the component of the fundamentalfrequency in the bridge voltage and the phase addition Δφ. The phaseaddition Δφ is chosen according to the same criteria as described abovewith reference to FIG. 7.

The signal-synthesizing member SSD, which is supplied with the valuesC_(a) and φ_(a) and the above-described cosine signal SCOS and the sinesignal SSIN, respectively, comprises a calculating member CALC4, adaptedto form as output signal an expression C_(a) cos(φa) and a calculatingmember CALC5, adapted to form as output signal an expression -C_(a)sin(φa). The output signal from CALC4 is multiplied by the cosine signalSCOS in a multiplier 26 and the output signal from CALC5 by the sinesignal SSIN in a multiplier 27. The products thus obtained are summed ina summator 28, the output signal of which forms the damping signal UOD.

In this case, thus, the damping signal may be written asUOD=DCOS*cos(2πf₀ t)+DSIN*sin(2πf₀ t), where the amplitudes DCOS andDSIN are formed in dependence on the product of the output signal IOCfrom the function-forming member 16 and the amplitude C₁ of thecomponent of fundamental frequency in the bridge voltage and of thephase addition Δφ.

Also in this embodiment of the invention, the damping signal is suppliedto the selector SEL2, which for sake of clarity is also shown in FIG. 9.In this embodiment, the limiting signal AMAXLI is formed in a manneranalogous to that described above with reference to FIGS. 6 and 7, butwith the difference that the selector SEL3 in this case is supplied withthe compensating amplitude signal C_(a) (FIG. 9).

FIG. 12 shows a method of forming the detector signal FFD, which may beused to advantage in an embodiment of the invention as described withreference to FIG. 7. A detector circuit DET3 comprises a comparingmember 29, a releasing-delaying delay member 30 with the time delay t4and an operation-delaying delay member 31 with the time delay t5, whichare mutually cascade-connected in the order mentioned. The comparingmember is supplied with the amplitude signal AMPL and forms an outputsignal if the amplitude signal exceeds an optional threshold value CR.The time t5 may advantageously be chosen to be of the order of magnitudeof 40 ms and the time t4 may be chosen to be of the order of magnitudeof 50 ms.

FIG. 13 shows a method of forming the detector signal FFD, which may beused to advantage in an embodiment of the invention as described withreference to FIG. 9.

If the cosine amplitude signal A and the sine amplitude signal B,considered as components of a vector corresponding to the amplitudesignal AMPL, are conceived to be reproduced in a frequency plane, theresultant vector, when the measured value UDB of the bridge voltagecontains a component of a frequency which deviates from the fundamentalfrequency, will rotate at an angular frequency which corresponds to thedifference in frequency between the component in the bridge voltage andthe fundamental frequency. When the bridge voltage contains a componentof a frequency which is equal to the fundamental frequency, the vectorwill remain stationary in the vector diagram. By detecting whether thecosine amplitude signal A and the sine amplitude signal B, respectively,exceed an optional threshold value during an optional period of time,the band width of the detection, viewed in the frequency plane, will bedependent on the optional time.

The cosine amplitude signal A is supplied to a first detector circuitPET1 comprising a comparing member 321, which forms an indicating signalAOSC1 if the cosine amplitude signal A exceeds an optional thresholdvalue AR. The indicating signal AOSC1 is supplied to a sub-detectorcircuit DET1P, which comprises a first delay member 33 with a time delayt1 and a second delay member 34 with a negating input and a time delayt2. The output signal from the first delay member is supplied to the Sinput of a bistable circuit 35 and a third delay member 36 with the timedelay t3. The output signal from the second delay member is supplied toan AND circuit 37 and the output signal from the third delay member issupplied to a negative input of this AND circuit. The output signal fromthe AND signal is supplied to the R input of the bistable circuit forrestoring its Q output. If after time t1 the indicating signal AOSC1remains, the Q output is set at the bistable circuit. When theindicating signal AOSC1 disappears and remains gone for at least thetime t2, the Q output is restored after time t3-t2. However, the Qoutput always remains set at least for the time t3. Advantageous choicesof the respective delays have proved to be t1=40 ms, t2=10 ms, and t3=50ms.

The cosine amplitude signal A is also supplied to a comparing member 322which is comprised in the detector circuit DET1 and which forms anindicating signal AOSC2 if the cosine amplitude signal A is lower thanthe threshold value AR with a negative sign. The indicating signal AOSC2is supplied to a detector circuit DET1N, which is built up in a mannersimilar to that of the sub-detector circuit DET1P.

The sine amplitude signal B is supplied to a second detector circuitDET2 with a function completely analogous to the one described above.The output signals from the respective Q outputs in the sub-detectorcircuits are supplied to an OR circuit 38, the output signal of whichforms the detector signal FFD when the Q output from at least one of thebistable circuits of the sub-detector circuits is set.

The invention is not limited to the embodiments shown but a plurality ofmodifications are feasible within the scope of the inventive concept.Thus, in the event that a converter comprises more than one 6-pulsebridge, the above-mentioned compensating signal and the above-mentioneddamping signal are formed for each of the converter bridges and, in amanner known to the skilled person, a selector may be adapted to selectand supply to the control equipment of the converter the greatest of theabove-mentioned signals.

The adaptation of the amplification to the current level in theconverter, aimed at with the amplitude-adapting member AAD, may also beachieved by alternatively supplying the function-forming member 16 to asensed and low-pass filtered value ID of the direct current Id in theconverter.

The blocks shown in the block diagrams may in applicable parts bedesigned as a model comprising analog and/or digital means formodelling, or be completely or partially performed as calculations bymeans of analog and/or digital technique in hardwired circuits, or beimplemented as programs in a microprocessor.

By a method and a device according to the invention, the advantages ofthe series compensation may be utilized without the risk of commutatingerrors in connection with transients in direct current and/or controlangle increasing. For the inverter this is of importance since thecontrol equipment for this normally strives to reduce the commutatingmargin to a minimum. For the rectifier this is of importance above allin connection with rapid control-angle changes in connection with, forexample, a fault condition in the d.c. connection when the rectifier iscontrolled towards the highest possible negative voltage by ordering thecontrol angle to a value near 180°. The result of these control-anglechanges is, on the one hand, that the risk of unbalance of the mentionedkind increases and, on the other hand, that the rectifier will operatenear its commutating margin.

During faultless operation of the converters the damping signalcontributes to a faster damping of that component of the fundamentalfrequency in the direct voltage across the converter bridges which iscaused by unbalance voltages in the series capacitors.

We claim:
 1. A method for controlling a series-compensated converterstation included in an installation for transmission of high-voltagedirect current, said converter station comprising a converter (SR1,SR2), controlled by control equipment (CE1, CE2), with at least one6-pulse bridge (BR), said 6-pulse bridge via series capacitors (SCR,SCS, SCT) being connected to a three-phase alternating-voltage network(N1, N2) with a fundamental frequency (f₀₁, f₀₂), said control equipmentgenerating an ordered value (AOL) of a control angle (α) for valves(V1-V6) included in the 6-pulse bridge in dependence on a limitingsignal (AMAXL) capable of being controlled, said method comprising thesteps of: forming an amplitude signal (AMPL) which corresponds to theamplitude (C1) for a component (C1^(cos) (2πf₀ t+φ1)) of the fundamentalfrequency in the direct voltage (Udb) of the 6-pulse bridge,continuously calculating a compensating signal (ACOMP) in dependence ona sensed voltage (Un1, Un2) at the converter station and on theamplitude signal according to a relationship (H0, H1) which at leastapproximately imitates a relationship according to which a control-angleaddition (Δα), at a commutating margin (τ_(m)) for the valves equal to apreselected value (τ_(p)), is a function (F0) of a current (Id1, Id2) inthe converter station, said voltage in the converter station and theamplitude (C₁) for said component of the fundamental frequency in thedirect voltage of the 6-pulse bridge, and forming the limiting signal independence on the compensating signal for the purpose of maintaining thecommutating margin (τ_(m)) for the valves equal to at least thepreselected value (τ_(p)) in case of unbalance between the voltages(UCR, UCS, UCT) of the series capacitors.
 2. A method according to claim1, wherein the compensating signal is calculated in dependence on avalue (UDI0) of the ideal no-load direct voltage of the 6-pulse bridgeand on the amplitude signal according to a relationship ##EQU5## whereAMPL is the amplitude signal amplitude signal and K1 is an amplificationfactor.
 3. A method according to claim 1, wherein a damping signal (UOD)is formed in dependence on said component of fundamental frequency inthe direct voltage of the 6-pulse bridge, and whereby the phase angle(φ_(a)) of the damping signal is formed as a difference (φ₁ -Δφ) ofphase angle (φ₁) for said component of the fundamental frequency and aphase addition (Δφ), and wherein the ordered value (AOL) of the controlangle for the valves included in the 6-pulse bridge is generated independence on the damping signal.
 4. A method according to claim 3wherein the amplitude of the damping signal is formed in dependence on aproduct of the amplitude (C₁) for said component of fundamentalfrequency and of a value (IOC) which is formed in dependence on acurrent level (IO, Id) for the converter and which decreases withincreasing current level.
 5. A method according to claim 1 wherein adetector signal (FFD) indicating the occurrence of said component offundamental frequency is formed according to a predetermined criterionand wherein the limiting signal is formed in dependence on thecompensating signal and on the detector signal.
 6. A method according toclaim 3, wherein a detector signal (FFD) indicating the occurrence ofsaid component of fundamental frequency is formed according to apredetermined criterion and wherein the ordered value of the controlangle for the valves included in the 6-pulse bridge is generated independence on the damping signal and on the detector signal.
 7. A devicefor control of a series-compensated converter station included in aninstallation for transmission of high-voltage direct current, saidconverter station comprising a converter (SR1, SR2), controlled bycontrol equipment (CE1, CE2), with at least one 6-pulse bridge (BR),said 6-pulse bridge via series capacitors (SCR, SCS, SCT) beingconnected to a three-phase alternating-voltage network (N1, N2) with afundamental frequency (f₀₁, f₀₂), said control equipment generating anordered value (AOL) of a control angle (α) for valves (V1-V6) includedin the 6-pulse bridge in dependence on a limiting signal (AMAXL) capableof being controlled, and voltage measuring devices (UM1, UM2) forsensing the direct voltage (Udb11, Udb12, Udb21, Udb22) of at least oneof the 6-pulse bridges, wherein the control equipment comprises anamplitude-value forming unit (AVU), which is supplied with a measuredvalue (UDM) of the direct voltage of at least one of the 6-pulse bridgesand forms an amplitude signal (AMPL) corresponding to the amplitude (C₁)for a component of the fundamental frequency in the direct voltage (Udb)of the 6-pulse bridge, a calculating unit (CALC2), which is suppliedwith said amplitude signal and a value (UDI01, UDI02) of the idealno-load direct voltage (Udi01, Udi02) and forms a compensating signal(ACOMP) in dependence on a continuously calculated value of acontrol-angle addition (Δφ), calculated in dependence on a voltage (Un1,Un2) at the converter station and on the amplitude signal according to arelationship (H0, H1) which at least approximately imitates arelationship according to which the control-angle addition, at acommutating margin (τ_(m)) for the valves equal to a preselected value(τ_(p)), is a function (F0) of a current (Id1, Id2) in the converterstation, said voltage at the converter station and the amplitude (C₁)for said component of fundamental frequency in the direct voltage of the6-pulse bridge, and wherein the limiting signal is formed in dependenceon the compensating signal for the purpose of maintaining thecommutating margin (τ_(m)) for the valves equal to at least thepreselected value (τ_(p)) in case of unbalance between the voltages(UCR, UCS, UCT) of the series capacitors.
 8. A device according to claim7, wherein the calculating unit calculates the control-angle additionaccording to a relationship (H1) of the form ##EQU6## where AMPL is theamplitude signal and K1 is an amplification factor.
 9. A deviceaccording to claim 7 wherein the control equipment comprises adamping-signal forming unit (DSU), which is supplied with at least onesignal (C_(1m) cos(2πf₀ t+φ1), A, B, AMPL) corresponding to a componentof the fundamental frequency in the direct voltage of the 6-pulse bridgeand which forms a damping signal (UOD) in dependence on said at leastone supplied signal, the damping-signal forming unit comprising aphase-shifting member (PHS) which forms the phase angle (φa) of thedamping signal as a difference (φ1-Δφ) of the phase angle (φ1) for saidcomponent of fundamental frequency and a phase addition (φ1), andwherein the ordered value (AOL) of the control angle for the valvesincluded in the 6-pulse bridge is generated in dependence on the dampingsignal.
 10. A device according to claim 9, wherein the damping-signalforming unit is supplied with a value of a current level (IO, Id) forthe converter, and wherein the damping signal forming unit comprises anamplitude-adapting member (AAD) which forms the amplitude of the dampingsignal in dependence on a product of said at least one signal suppliedto the damping-signal forming unit and corresponding to a component ofthe fundamental frequency in the direct voltage of the 6-pulse bridgeand of a value (IOC) formed in dependence on said current level for theconverter, said value decreasing with increasing current level.
 11. Adevice according to claim 7 wherein the control equipment comprises adetector circuit (DET) which forms a detector signal (FFD) indicatingthe occurrence of said component of fundamental frequency according to apredetermined criterion, and wherein the limiting signal is formed independence on the compensating signal and on the detector signal.
 12. Adevice according to claim 9 wherein the control equipment comprises adetector circuit (DET) which forms a detector signal (FFD) indicatingthe occurrence of said component of fundamental frequency according to apredetermined criterion, and wherein the ordered value of the controlangle for the valves included in the 6-pulse bridge is generated independence on the damping signal and on the detector signal.